The present invention relates to an AC current control system for controlling an AC current such as that supplied to a load, e.g., an AC electric motor, by a power converter forming a variable voltage and variable frequency power source.
A variable speed control of an AC motor has been frequently used in which a power converter is used as a power source of an AC electric motor, and the voltage and the frequency are controlled to control the speed of the motor. Particularly, what is called a vector control has been devised in which a current reference which optimizes the response of the AC electric motor is calculated and the output current of the power converter, i.e., the current of the motor is controlled in accordance with the calculated current reference, thereby enabling a control of an AC electric motor with a quick response comparable with that in the control of a DC electric motor. To accomplish the vector control, it is necessary to have the current of the AC motor follow closely (i.e., with a high fidelity) the calculated current reference.
A cyclo-converter incorporating thyristors is known as a large capacity power converter. With a cyclo-converter, the firing of the thyristors must be conducted in synchronism with the phase of the AC power inputted to the cyclo-converter, and a delay is introduced in the switching control. For this delay, the output frequency becomes higher and a delay in phase (phase lag) and an error in the amplitude between the current reference and the actual current are caused.
For a small capacity power converter, a transistor inverter is used. With transistors, a quicker switching control is possible, so that the phase lag is relatively small. But, transistor inverters are used at a higher frequency, so that the phase lag and the amplitude error become problematical.
In a vector control, the current references i.sub.d *, i.sub.q * along two axes which are orthogonal or in quadrature with each other on a rotating coordinate are calculated and given to a current control device. The two-axis current references i.sub.d *, i.sub.q * may be considered to correspond or form counterparts of a field current reference and an armature current reference in a control of a DC electric motor, and are given in the form of a DC signal or a quasi-DC signal (consisting of a DC component and/or very low frequency component).
FIG. 1 shows a conventional current control system in which two-axis current references are used. As illustrate the current references i.sub.d *, i.sub.q * are transformed, by a coordinate transformer 1 into three -phase, i.e., U-phase, V-phase, W-phase, current references i.sub.u *, i.sub.v *, i.sub.w * which are three-phase sinusoidal signals having an angular frequency .omega. of the desired AC current. The AC currents of the respective phases are controlled in accordance with the references i.sub.u *, i.sub.v *, i.sub.w *. The U-phase current reference i.sub.u * is given by the following expression: EQU i.sub.u *=i.sub.d * cos .omega.t+i.sub.q * sin .omega.t (1)
The V-phase current reference i.sub.v * and the W-phase current reference i.sub.w * are values having a phase lagging by 2.pi./3 and 4.pi./3, respectively.
Actual load currents i.sub.u, i.sub.v, i.sub.w are detected by current detectors 10, 11, 12 and are compared at the subtractor 2, 3, 4 with the current references i.sub.u *, i.sub.v *, i.sub.w *. The resultant deviations are passed through control amplifiers 5, 6, 7, whose outputs e.sub.u *, e.sub.v *, e.sub.w * are supplied as voltage references to a power converter 9 converting a power from an AC power source 8 and supplying the converted power to a load in the form of an AC electric motor 13. The power converter 9 is thereby controlled to output voltages in accordance with the voltage references so that the actual load currents are made and maintained equal to the three-phase current references. In general, the control amplifiers 5, 6, 7 are in the form of P-I (proportional plus integral) control amplifiers incorporating a proportional element and an integration element. The gain frequency characteristic .vertline.G(.omega.).vertline. of each control amplifier is given by the following equation (2). ##EQU1## where K.sub.P represents a proportion gain, and
K.sub.I represents an integration gain.
For a component of the angular frequency .omega. being zero, i.e., the DC component, the gain is the infinity so that no steady-state deviation occurs. For components of larger angular frequencies .omega., the overall gain becomes closer to the proportion gain K.sub.P and the steady-state deviation appears in the form of a phase lag and an amplitude error. If the gains K.sub.P, K.sub.I are set at large values the steady-state deviation can be made smaller. But the switching control in the power converter has a delay and because of the delay the current control becomes unstable if the gains K.sub.P, K.sub.I are set at excessively large values.
To avoid the phase lag and the amplitude error, a current control system as shown in FIG. 2 was contemplated. In this system, three-phase current values i.sub.u, i.sub.v, i.sub.w in a stationary coordinate are transformed by a coordinate transformer 19 into orthogonal two-axis current values i.sub.d, i.sub.q in a rotating coordinate rotating at an angular frequency .omega., thereby forming a quasi-DC signals. The values i.sub.d, i.sub.q are compared at subtractors 14, 15 with two-axis current reference values i.sub.d *, i.sub.q * and the differences are passed through control amplifiers 16, 17, whose outputs forming two-axis voltage reference values e.sub.d *, e.sub.q * area transformed by a coordinate transformer 18 into threephase voltage reference values e.sub.u *, e.sub.v *, e.sub.w *, and are supplied to a power converter 9. The coordinate transformer 18 is similar to the coordinate transformer 1 and the relationship between the inputs and the outputs is similar to that of the equation (1). The relationship between the inputs (the three-phase current values i.sub.u, i.sub.v, i.sub.w) and the outputs (the two-axis current values i.sub.d, i.sub.q) of the coordinate transformer 19 is shown by the following equations: ##EQU2##
If the amplitude of the three-phase currents is I, and the phase of the U-phase current is represented by .theta. (the phases of the V-phase current and the W-phase current are lagging by 2.pi./3, 4.pi./3 respectively), the three-phase currents are given by the following equations: EQU i.sub.u =I cos (.omega.t+.theta.) (5) EQU i.sub.v =I cos (.omega.t+.theta.-2.pi./3) (6) EQU i.sub.w =I cos (.omega.t+.theta.-4.pi./3) (7)
Substituting i.sub.u, i.sub.v, i.sub.w in these equations for those in the equations (3), (4), we obtain: EQU i.sub.d =I cos .theta. (8) EQU i.sub.q =I sin .theta. (9)
It is seen that i.sub.d, i.sub.q are quasi-DC values which are independent of the angular frequency .omega.. Thus, the coordinate transformer 19 transforms AC signals in a stationary coordinate into quasi-DC signals in a rotating coordinate, while the coordinate transformer 18 transforms quasi-DC signals in a rotating coordinate into AC signals in a stationary coordinate.
It was expected that since the quasi-DC, two-axis current values i.sub.d, i.sub.q are compared with quasi-DC, two-axis current reference values, effect of the angular frequency .omega. would be eliminated and the phase lagging and amplitude error would be detected as a DC component, and DC component control by means of the integrating element of the control amplifier would reduce to zero the phase lagging and the amplitude error. However, it has been found that stable control is difficult to achieve. The reason for this is, it is considered, that the control is, in essence, not a true DC control but a control using quasi-DC signals derived by transformation from AC signals.
As an improvement, Japanese Patent Application Laid-open (Kokai) No. 52392/1982 discloses a current control system in which the conventional direct control of the three-phase currents and the abovedescribed quasi-DC control are combined. This is shown in FIG. 3, in which blocks 1-7 are the same as those in FIG. 1, and blocks 14-19 are the same as those in FIG. 2. In addition, adders 20, 21, 22 are provided to add the outputs of the control amplifiers 5, 6, 7 forming part of the conventional three-phase current control and the outputs of the coordinate transformer 18 forming part of the quasi-DC control by means of coordinate transformation. It was reported that by the combination of both control features, phase lagging and amplitude error can be eliminated (1980 National Convention Record of Electrical Engineers of Japan, No. 504; published Apr. 2, 1980). But, as will be readily seen, the complicated control circuitry is required. Moreover, to achieve the three-phase current control, which is an AC control, by digital means, such as a microcomputer, the sampling and calculation must be repeated at a rate corresponding to the maximum angular frequency .omega..sub.M. Generally, the sampling frequency must be 10 times the maximum frequency (.omega..sub.M /2.pi.) of the AC current. If the maximum frequency of the AC current is 100 Hz, the sampling frequency must be in the order of 1 KHz. Such highrate sampling and calculation are practically impossible when a microcomputer is used. In a DC control or a quasi-DC control, the sampling frequency need only be 10 times the maximum frequency of the rate of change of the DC current or the two-axis current values i.sub.d, i.sub.q. The maximum frequency of the rate of change is generally in the order of 10 Hz, so that the sampling frequency need only be in the order of 100 Hz. As a result, use of a microcomputer for the control amplifiers 16, 17 in FIG. 3 is easy. But even if these circuits are realized by a microcomputer, the circuits 5, 6, 7 for the three-phase current control must be formed of analog circuits or special purpose digital circuits. Thus, simplification of the hardware, improvement in reliability and cost-down of the device were not fully accomplished.